Readers, please note this page is presented for your education, information and guidance only - it is not intended to be a technical/scientific treatise. The concepts and ideas presented herein are just as much subjective as objective.

It attempts to assist those experienced audiophile home constructors who wish to explore further tube amplifier development options and are prepared "to go beyond the square" - to challenge the paradigm of "the status-quo".

This paper refers only to the characteristics and performance of push-pull tube audio amplifiers without negative feedback.

For reasons detailed elsewhere in my website I have no interest whatsoever in single-ended amplifiers - however the concepts described herein are as equally applicable to single-ended amplifiers as to push-pull configurations.

For reasons detailed elsewhere in my website I have no interest whatsoever in the use of trans-stage negative feedback thus the concepts described herein are intended to optimise electron tube performance without the use of negative feedback.

For full ratings and applications of specific tube types in which you are interested please refer to the manufacturer's catalogue.

Copyright in all quoted works remains with their owner, author and publisher, as applicable.

Please note that no warranty is expressed or implied - see footnote notice.

Excepting where rights in Intellectual Property previously exist, all rights in the applied engineering concepts expressed in this paper remains exclusively with the author.

The term OPTIMISED ELECTRON STREAM TECHNOLOGY applies to all part or whole uses of the concept.

The whole or part thereof of this paper and/or the designs and design concepts expressed therein may be reproduced for personal use only without limitation - but must not under any circumstances be applied or used for commercial gain or reward without the express written permission of the author, being the intellectual copyright owner.

All rights reserved.




It is recommended that to best understand the concepts presented here, the reader first carefully study my explanatory papers.

To directly access please click on each of the links below:







It is intended that the set of OPTIMISED ELECTRON STREAM TECHNOLOGY concepts presented below represent - individually or together as a set - an innovative extension of audio engineering design thinking to the above linked presentations elsewhere on my website, and a major breakthrough in electron power tube technology available to the home audio constructor.

Historically we have been restricted by convention to a small set of engineering design principles that have imposed a barrier to further development, reinforced by commercial manufacturers staying with the tried and true "safe" design configurations.

However the availability of low-cost high-performance solid state amplifiers and pre-amplifiers, together with advances in recording formats - currently Super Audio CD and DVD standards - have provided stimulus to tube hi-fi enthusiasts to improve the performance of their existing equipment.

Many of the old constraints are longer with us - hiss, noise, wow, flutter and rumble etc. do not present on CD or DVD - so enable us to open the frequency range and dynamic range window a little more.

Modern CD's and DVD's offer a substantially higher dynamic range than traditional analogue vynil recordings. This dynamic range is compressed when the amplifier is not capable of reproducing it, resulting in a loss of fidelity and realism.

The OBJECT of OPTIMISED ELECTRON STREAM TECHNOLOGY is to OPTIMISE performance in a high-fidelity audio amplifier by simultaneously loading ALL the internal elements of a power electron tube with a load directly proportional to their ACTUAL applied voltages - AC and DC - as distributed between the positive and negative terminals of the tube - ie anode and cathode - AND in direct proportion to their physical electrode spacing (dielectric strength).


OES  TECHNOLOGY offers superior performance because it is simple and because the tube is always in equilibrium.

The conceptual model is illustrated in the following circuit diagram for Beam Power Tubes, Tetrodes and Pentodes in conventional or Ultra-linear mode  - as applicable:

To prepare a foundation for the concepts presented on this page, please read my SCREEN-GRIDS page before reading this paper, because the function and behaviour of the Screen-grid is fundamental to OPTIMISED ELECTRON STREAM TECHNOLOGY.

In the case of ultra-linear connected TETRODES and BEAM POWER TUBES only the Screen Grid is connected to the 40-50% turns transformer primary winding tap as shown above (because there is no separate Suppressor Grid terminal on the tube)


This paper introduces nine new applied engineering OPTIMISED ELECTRON STREAM TECHNOLOGY concepts for refining and optimising the electron flow in a vacuum tube and its control circuitry.

Subject to individual circuit design, all or any of the above OPTIMISED ELECTRON STREAM TECHNOLOGY concepts may be used individually, or in combination, or all together - each offering a specific benefit to enhancing electron flow within each tube or circuit.


       OPTIMISED SCREEN-GRID VOLTAGE AND LOAD for Tetrodes, Pentodes and Beam Power Tubes in "Ultra-linear", or "Distributed Load" operation.
    OPTIMISED SCREEN-GRID VOLTAGE AND LOAD for Tetrodes, Pentodes and Beam Power Tubes in "Pentode' operation
2.1     OPTIMISED SCREEN-GRID VOLTAGE AND LOAD for Tetrodes, Pentodes and Beam Power Tubes in "Ultra-linear", or "Distributed Load" operation.
My OPTIMISED ULTRA LINEAR   page describes a design method for optimising the DC supply voltage and load impedance to the Screen Grids in Tetrodes, Pentodes and Beam Power Tubes for "Ultra-linear", or "Distributed Load" operation.
To determine what optimum transformer primary tap ratio to use it is essential to physically determine the actual Screen Grid to Anode physical (linear distance) space ratio, because for general audio class electron tubes the ratio of screen grid to plate voltage will be optimally equal to the physical distance ratio of cathode to screen-grid v cathode to plate.
2.2      OPTIMISED SCREEN-GRID VOLTAGE AND LOAD for Tetrodes, Pentodes and Beam Power Tubes in conventional "Pentode' operation
The same principle for optimisation of tube performance may be applied to conventional operation of Tetrodes, Pentodes and Beam Power Tubes in "Pentode' operation - ie, where the Screen-Grids are supplied from a separate DC source to the Plates.

To optimise the Screen-Grid voltage, first carefully and accurately measure the physical gap spacing between the Cathode, Screen-Grid and Plate.

In most cases this can be easily done without having to destroy a valuable tube. Just measure the spacing between Cathode and
Anode and of the grid support pins at the top of the tube. In the case of beam power tubes this may be difficult because there is
usually a box-like assembly covering the Plate structure. But a bit of ingenuity should solve the problem

If it becomes necessary to destroy a tube to examine its internals safety precautions must be observed:

1.    Wrap the tube in a strong cloth
2.    Place the tube in a vyce and squeeze the glass bottle slowly until it implodes
3.    Gently hit the glass with a hammer at a point not directly over the electrode assembly
4.    Carefully remove the broken glass and dispose of safely
5.    Carefully cut away the electrodes until the elements can be measured

The "gap" spacing ratio can then be calculated for the particular tube type.

To determine the Optimised Screen-Grid voltage, multiply the Plate voltage by the gap ratio.

It is obvious the Screen-Grid voltage will be substantially less than the Plate voltage - typically in the range 30 to 50 % but usually 40% - so a separate stable supply is essential.

The designer has the option of using:

What is most important is for the Screen-Grid DC supply voltage to be reasonably constant between no-signal load and full-signal load conditions (ie well regulated).

This can be effectively achieved by using a separate power transformer of generous rating - ie at least twice the maximum signal DC Screen-Grid current, a full-wave silicon bridge rectifier and humungous filter capacitors in capacitor input to filter configuration - preferably with a low DC resistance filter choke installed too.

For the more theoretical designer some thought could be given to the "Virtual Cathode" concept, which suggests that the portion of  electron stream devoted to the negative bias applied to Grid #1 creates a more negative "Virtual Cathode" in the region of the physical Grid element. This concept however is difficult to assess because so long as the tube is conducting some of the electron stream is continuous between real Cathode and Plate. Maybe only the outer portions of the electron stream are affected by the negative bias and not the entire thickness of stream.

The perfectionist - or those having the necessary equipment - looking for the ultimate optimisation could determine the optimum operating voltage for the Screen-Grids by means of a Signal Generator, Distortion Meter and variable voltage power supply to the Screen-Grids.

The optimised AC load impedance for the Screen-grid will be the ratio of DC Screen-grid volts to Plate volts x Plate load impedance.
If the Screen-grid load is provided by a fixed resistor the ratio for a Screen-grid - or set of Screen-grids in one side of a push-pull set, will be the DC voltage ratio x one quarter Plate to Plate load impedance

Note however that if the load is provided by a transformer as in ultra-linear or equivalent mode, the Screen to Screen load impedance will be directly proportional, but the turns ratio will be the square of the DC voltage ratio.

eg If the DC ratio is 40% then the turns ratio for the Screen-grid tap on the output transformer will be 0.4 x 0.4 = 0.16 or 16%

A technical paper describing benefits of reduced screen-grid voltage is presented at http://www.tubebooks.org/Books/lockhart.pdf

Recent Research

I am indebted to Rudolf  Moers, a distinguished Electrical and Electronics Engineer located in the Netherlands, who has made available for us his wonderful recent scientific investigation into the design theory and practice of Ultra-linear audio amplification.

These papers are posted with permission from Linear Audio www.linearaudio.net and their author Rudolf  Moers.

Part 1 - Power Point presentation

Part 2 - Power Point presentation

Parts 1 and 2 combined presentation - pdf format

Paper - The Ultra-Linear Power Amplifier: An adventure between triode and pentode - pdf format

The engineering design methods developed by Mr Moers may be used to determine theoretical plate/screen load ratios for ultra-linear operation of power tubes.



The PENTODE electron tube has an extra grid (Grid #3) installed to suppress the effects of secondary emission, as described in my SCREEN-GRIDS page.

Although few true Pentodes are available for audio power amplifier output stage applications, this section relates to them because they are suitable for use in OPTIMISED ELECTRON STREAM TECHNOLOGY applications.

Manufacturers' tube data sheets generally describe Pentodes as having higher measured distortion than Beam Power Tubes so most audio amplifier designs have focused on the latter types.

In the case of recycled or salvaged tubes from yesteryear, it is evident that by WWII, RF design engineers preferred Beam Power Tubes because of their capability to operate at higher RF frequencies, hence the true Pentode enjoyed only a short claim to fame.

One exception to this is the mighty 803 Pentode, which found widespread use in submarine and destroyer RF applications because of its ruggedness, stability, reliability and long-life. Of interest is that the 803 pentode has aligned grids, each having a ceramic coating to limit electron attraction, thereby improving predictability and stability of operation, as well as efficiency.

The 837 Pentode tube is also renown as a very reliable oscillator tube for RF transmitters. The 837 Pentode has a Plate structure similar to that of the famous TT21 and KT88 Beam Power Tubes.

However Pentodes do offer user benefits over Tetrodes and have a place in high-fidelity audio and RF amplifiers - particularly for home constructors who might have a box of useful Pentodes just waiting to be used in a suitable design.

It is a well established tube engineering principle that the current in a tube can be regulated using ANY of the grid elements.

For example, in domestic radio receivers, the use of multi-grid tubes such as 6A7, 6A8 and 6L7, each having 5 grids, is a standard application of radio engineering design.

However for audio engineering purposes, three grids appears to be the practical limit - beyond which no appreciable benefit is to be realised. In fact. the beam Power Tube utilises only two grids to control current through the tube.

Using the same applied engineering technology as developed for OPTIMISED ULTRA LINEAR operation, it is possible and practicable in power tubes fitted with a SUPPRESSOR-GRID having an independent base pin connection, to apply a DC voltage and AC load impedance to the SUPPRESSOR-GRID having the same ratio to Plate voltage and load respectively, as the internal electrode physical gap ratio.

IMPORTANT NOTE 1: It is the case in many of the popular PENTODES that the SUPPRESSOR-GRID is internally connected during manufacture to the CATHODE, so it is not possible to externally access the Suppresser Grid. In this class of tube the manufacturer controls the behaviour of the Suppressor-grid and the user cannot do anything to change that.

IMPORTANT NOTE 2: Tube identification protocols describe and illustrate by base pinout diagrams Beam Power Tubes as "Pentodes". These tubes are essentially Tetrodes that have a beam forming electrode used to both confine the electron stream to a pre-determined width within the tube, and to focus the electron flow onto a particular area of the Plate. They usually have aligned Grids to arrange the electron flow in multiple sheets.

Therefore it is essential to physically examine tubes of interest to determine their actual mechanical construction.


However those tubes having a separate base pin connection for the SUPPRESSOR-GRID offer the designer the option of applying a suitable DC voltage consistent with the proportional voltage divider effect of the internal electrode gap.

The rules and requirements for an adequately regulated DC power source are the same as described above for the Screen-Grid.

For example, in the Type 803 - 125W dissipation power transmitting pentode tube - the Suppressor Grid is set at nominally 68% of the Cathode to Anode gap. Therefore the OPTIMISED ELECTRON STREAM value for the Suppressor Grid DC voltage will be the applied Plate Voltage x 68% - but only up to a value 158% of that permitted by the manufacturer's rated Screen-Grid Voltage (because in the case of Tube Type 803, the maximum permissible Screen-Grid voltage is nominally 43% of the applied Plate Voltage).

In other words, the maximum applied DC Screen-Grid Voltage and/or Suppressor Grid Voltage for OPTIMISED ELECTRON STREAM operating conditions will be determined by either the maximum rated Screen Grid or Suppressor Grid voltage - whichever is the lesser.


The SUPPRESSOR GRID (GRID #3) regulates the flow of electrons in the tube in the same way as is the case for Control-grid #1 and Screen-Grid #2.

In conventional "Pentode" configuration, the Suppressor-grid is directly connected to the Cathode either internally by the tube manufacturer or by the user.

Having left the Plate as surplus, randomly travelling electrons, they find their way to the Suppressor-grid, thence diverted to the Cathode to be absorbed back into the electron stream. This arrangement obviously creates a short-circuit in respect of those electrons attracted to the Suppressor-grid.

Thus there is an effective internal or external circuit (as applicable) created between the Suppressor-grid and the Cathode, that diverts some of the electrons back to the Cathode. This current is lost to the output power and therefore reduces efficiency in the output power stage.

In the case of the relationship between the Screen-grid and Plate, most experts suggest that the Plate sees the Screen-grid as the "Cathode", thus if this is so then the DC potential between Plate and Suppressor Grid will be again determined by the linear distance between them - unless the Suppressor-grid is purposefully connected to the Cathode.

It follows that in an OPTIMISED ELECTRON STREAM technology amplifier where the Screen-grid DC voltage is around 50% of the Plate voltage, if the applied DC voltage is increased from 0 volts (Cathode potential) to a positive voltage greater than the Screen-grid voltage, then the free electrons deflected from the Plate will still be attracted to the Suppressor-grid because it is still negative to the Plate.

It also follows that because the Suppressor-grid is now positive to the Screen-grid, Plate current will increase - albeit slightly.

Since both Screen-grid and Suppressor grids DC voltages will be fixed, it becomes obvious that to control the electron flow within permissible limits, the negative bias voltage applied to Grid #1 Control Grid will need to be made MORE NEGATIVE.

It is also obvious that to limit the DC current flow in Grid #3, and to prevent an AC signal short-circuit at the Suppressor-Grids,  it is essential to load the Suppressor-grids by installing a Grid-stopper resistor of around 75% of equivalent Plate to Plate push-pull load impedance.

eg for a Plate to Plate load of 8,000 Ohms, the transformer will present a load of 2,000 Ohms to each tube in the push-pull pair.
2,000 Ohms x 75% is 1500 Ohms. This is still a relatively small value so should not present significant voltage drop or regulation issues.

This is an important difference between convention and the OPTIMISED ELECTRON STREAM configuration.

In a conventional pentode circuit with grounded Cathodes, the Suppressor-grid is directly connected to the Cathode so the Suppressor-grid is fixed at 0 VDC. However in the case of  OPTIMISED ELECTRON STREAM configuration, the Suppressor-grid will be fixed at a relatively high DC voltage set to enable the AC voltage to be aligned with the physical gap in the tube. The circuit design then needs to accommodate this to protect the Suppressor-grid from self-destruction through over-current.

The optimised AC load impedance for the Suppressor-grid will be the ratio of DC Suppressor-grid volts to Plate volts x Plate load impedance.

If the Suppressor-grid load is provided by a fixed resistor the ratio for a Suppressor-grid - or set of Suppressor-grids in one side of a push-pull set, will be the DC voltage ratio x one quarter Plate to Plate load impedance

When used in an OPTIMISED ULTRA LINEAR configuration as shown above, it will also will be loaded by the tapped primary winding on the output transformer. The AC load impedance will be determined by the ratio of the physical gap between the Cathode and Suppressor-grid compared to the Cathode to Plate, as applicable - nominally 75% of the Plate to Plate load.

The Suppressor-grid to Suppressor-grid load impedance will be directly proportional, but the turns ratio will be the square of the DC voltage ratio.

eg If the DC ratio is 75% then the turns ratio for the Suppressor-grid tap on the output transformer will be 0.75 x 0.75 = 0.5625 or 56.25%

Consequently, proportionate audio POWER is drawn off from the Screen-grid and Suppressor-grid by the output transformer.

PENTODE tubes suitable for high-fidelity audio power output applications using OPTIMISED ELECTRON STREAM technology include:


I have not listed the EL34/6CA7, its smaller brother the EL84/6BQ5 and cousin 6M5,  because the construction and effect of the Suppressor-grid in these tubes is "nominal" and not in the same league as those of the transmitting tubes listed above. Being manufactured from very fine wire, the Suppressor Grid in these tubes is not capable of handling significant current or power. For high-powered audio amplifiers superior options are available as shown above.

However the EL34/6CA7,  EL84/6BQ5 and 6M5 are suitable for effective use in OPTIMISED ELECTRON STREAM amplifiers where the Screen-grid is set at 40% of Plate voltage and Suppressor-grid is connected directly to the Cathode in the usual way, and in OPTIMISED ULTRA LINEAR amplifier designs.


Consider also OPTIMISED ELECTRON STREAM TECHNOLOGY - SILICON DIODE FEED (as described in Section 4 below) to the Suppressor Grid.

In this case, a conventional pentode configured amplifier - where the pentode tube has a separate externally connected independent Grid 3 pin (typically connected to the cathode or ground) - may be modified by installing a silicon diode between the Suppressor-grid pin on the tube socket and Cathode or ground - as applicable. The arrow must point towards ground - ie the marked terminal on the diode to the Suppressor-grid pin.

This configuration allows AC current to flow to ground but blocks DC current into the tube.

If any instability or non-linearity occurs, a 1,000 Ohm resistor may be installed shunting the diode, to create a permanent DC current path to the tube and anchor the Suppressor Grid to the Cathode or ground.




Conventional historic tube amp design using tetrode, pentode, or beam power tubes, evolve about a push-pull output stage configuration typical of the sketch below:

In the example cathode-bias is shown, however the same principles apply for a fixed-bias design.

The notable feature of this design is that everything is symmetrical and so is easy to draw, read and understand.

The configuration of each output tube is considered to be equally distributed about the central axis of the output stage, (being the B+ and centre-tap of the output transformer).

The core electronic design concept assumes (deems) the Plate current in each tube will be identical at all times and therefore any distortion in the plate circuits is cancelled out via the push-pull effect. For an explanation of this refer to the RCA Tube Manuals.

Screen-grids are not considered to be more than a power output/efficiency enhancing benefit derived from tetrodes v triodes, and as shown in the example above, are not connected in any manner other than the most basic available - ie connected together and supplied from the same power source as the Plates.

So in theory, all of the components operate in a synchronous manner such that the output stage is efficient and delivers its power at low distortion.

Unfortunately things are not always as they appear.

The standard tolerance on components is usually around + or - 10%, however some components, including the tubes themselves, are assigned much wider tolerances.

Modern resistors are typically + or - 5% however + or - 10% tolerance is still used.

Modern capacitors display tolerances that vary with construction material and style.

Electrolytic capacitors typically display -20% + 100% on nominal capacitance, thereby affecting gain across the frequency range.

Vacuum tubes offer a tolerance of + or - 20% on transconductance when new. Depending upon circuit parameters, performance usually varies downwards with use. If one power tube draws grid current early on then performance of the pair will suffer.

Transformers vary dramatically. Even if the primary centre-tap is exactly in the centre of the total number of turns - ie the number of turns either side of the centre-tap is exactly equal - the transformer may still not deliver an equal power transformation between primary and secondary if the magnetic properties of the two halves of the primary are not identical. Also, if the secondaries are not equally distributed about the primary the induction into each secondary may vary between halves. Factors such as core design and winding design will influence the end result.

And what about the symmetry of the driver stage? This too is subject to tolerance variables and may deliver an asymmetrical signal drive voltage to the output tubes, resulting in uneven power output from each half.

So what appears to be a perfectly symmetrical design is in fact a widely varying practical configuration.

One obvious variant is Plate-current.

Many of the popular tube types do not deliver a linear response between zero signal conditions and maximum signal conditions. Whilst most guitar amp users would be familiar with the concept of "re-biasing" after changing an output tube, it is not the norm in hi-fi equipment. However despite good intent, guitar amp (and hi-fi amp) users are blissfully unaware that their beautifully balanced output stage at zero signal is nowhere near balanced at maximum signal.

In lower-cost/lower quality transformers it will be found by measurement that the DC resistance of one half of the primary is somewhat different to that of the other half - resulting in uneven Plate voltage as Plate-current increases.

Imbalance in a push-pull output stage results in loss of power and increased distortion.

This condition is more likely in high-gain tubes like the EL34/6CA7, EL36/6CM5, KT88 and 6550.


Ideally, the output from the Screen-grids is also identical and therefore cancels out at the output transformer centre-tap.

But note from the above design that any voltage appearing at the centre-tap of the output transformer - whether it be derived from a difference in Plate characteristics, transformer characteristics, untransformed signal, reflected back emf from the loudspeaker, or simply hum as a ripple voltage - will appear on one or the other Screen-gid as either direct injection or feedback.

Obviously Screen-grid current will normally be a portion of (ideally balanced) Cathode-current, but if different brand tubes - or tubes of differing design from the same manufacturer - are used in the output stage pair it is likely that the Screen-grids will behave differently - resulting in not only different tube characteristics across the range but also differences in the Screen-grid performances. Imbalance becomes apparent as distortion.

It follows then that a useful object of amplifier design would be to eliminate the Screen-grid as a variable from the system.

This proposition is very well explained by Renato D. Tancinco of the Philippines, in his 1961 US Patent 3153766

In this patent, Tancinco presents his design, which aims to eliminate crossover distortion in tetrodes and pentodes operating in Class B mode - EL34/6CA7 users take note.

Unfortunately this design only works in Class B mode because it cancels out part or whole of the opposite polarity alternating signal in the output stage, however it does offer significant advantage as the patent itself explains.

As noted in my Screen-Grids paper linked above, reference to tube handbooks shows that in a typical beam power tube, the Screen Grid current at maximum signal power is around 20% of Plate current. This ratio of currents appears to be largely independent of Plate voltage. It would therefore be reasonable to assume that up to 20% of prospective signal power is lost in the Screen Grid circuit in a conventional amplifier. (Power = volts x amps. Power supply voltage to the screen-grids can be up to 100% of plate voltage)

(Note: Two notable exceptions are the 807 and 814 beam power tubes that incorporate advanced design technologies to increase tube efficiency and reduce distortion, however in the overall scheme of things this technology appears to have been limited to these two tube types - if you are aware of others please let me know)

Two conventional options present to overcome this nominal ratio of 20% plate Current:

The first is ultra-linear connection, where all the electrons collected by the Screen Grids are fed into the output transformer, but in the process modify the output stage characteristics.

The second is to increase the value of the Screen Grid-stopper resistor to a value sufficiently high to resist the flow of electrons to the Screen Grid.

However the Grid-stopper resistor must be non-inductive to prevent oscillation. It must also be capable of handling the Screen current passing through it. One way of doing this is to use parallel carbon composition resistors (not film type - to prevent fire) of sufficient number to obtain the required heat dissipation rating to do the job without excessive temperature rise in the resistors.

A further problem here is regulation of the Screen Grid supply. Obviously a Grid-stopper resistor of say 5,000 - 10,000 ohms will present a significant voltage drop when Screen Grid current flows - if it does.

The loss of regulation may be a price we have to pay to obtain a high standard of performance.

Another and previously unpublished option to creating an operating environment where the Screen Grids will be at a DC potential sufficiently high enough to attract and accelerate electrons towards the Plates but - to maximise power output - not to collect and divert them to earth through the B+ supply, is the humble silicon diode semi-conductor rectifier.

By inserting a standard half-wave silicon rectifier diode in series with the Grid Stopper resistor, an electronic control circuit is created whereby the Screen Grid will be able to be energised at DC potential attracting and accelerating electrons towards the Plate - still electrostatically controlling current flow in the normal way - but blocking the flow of AC current from the Screen Grid back to the DC source - ie "one way traffic"

This works because the current flow in the tube is always from the Cathode to the Anode (Plate). The diode, being a semi-conductor, blocks current flow in the reverse direction, thus enabling DC current to feed it in the conventional manner but blocks AC current from passing back through it to a load.

Thus then there is no output circuit formed between the Screen-grid and the load so no current can flow in the usual direction.

Don't believe it ?

See https://www.youtube.com/watch?v=CdADlclVMZM 

A diode is a diode !!

This causes the tube to appear at first glance to behave like a triode. Note however that the tube is still a tetrode, pentode or beam power tube as applicable - it is not a "super triode" as some folks suggest. Plate-current is still controlled by Grid #1 and Grid #2 voltages in the usual way - the difference being that the Screen Grid signal current is not shorted to the B+ supply (AC ground) so is diverted to the Plate and collected as significantly additional power output.

The diode is connected between the B+ supply (line) and the Screen Grid (load) such that the arrow points towards the Screen Grid. ie forward current is from the line to the load.

This arrangement offers huge benefits, because it prevents the Screen Grids from collecting electrons - thereby diverting all the signal output to the Plates, increasing tube efficiency, reducing distortion and increasing frequency response, as well as eliminating the usual effects on changes in Screen Grid voltage on Plate Current - therefore improving transient response.

One major benefit is that the diode is not in the signal path and therefore does not modify the sound.

A secondary benefit is that there are no bypass or power supply capacitors (eg paper, polyester or electrolytic) in the Screen-Grid signal path, which is a further major improvement.

A further benefit is that the non-linearity described under Fig. 2 above will be less of a problem for us because the Screen-Grid component of the signal (ie those electrons normally collected by the Screen-Grid)  is diverted to the Plate.

Thus by inserting the humble silicon semi-conductor diode in the output circuit, we can completely break the bonds of traditional audio practice and take a great leap forward!!

This is not tube heresy, because the diode is not in the signal path - it merely prevents the signal from being affected by adverse circuit parameters such as short-circuited Screen-Grids, fluctuations in Screen-Grid voltage and power supply filter capacitors.

The relatively low reverse resistance of the diode appears to adequately satisfy the need for a low impedance return path to AC earth, so the Screen Grid is not actually isolated from AC earth - but there is sufficient impedance in the circuit to discourage electron flow through it.

Note: Having regard to the EIMAC articles regarding secondary emission in tetrodes referenced above, it should be the case that the use of a silicon diode in the Screen-Grid supply will not impede reverse current flow - provided a suitable bleeder resistor is used between the diode and B+ source - as would be the case without the diode in the circuit.

Some audiophile experimenters have used single or series strings of zener diodes in this kind of circuitry to regulate the DC Screen Grid voltage derived either directly from the Plate or from the B+ supply, however I have used single conventional 1A 1000 PIV silicon rectifiers (from B+ supply only) with good results.

Important Note: When Zener Diodes are inserted in series with the Screen-Grid supply to both drop and control Screen-Grid voltage, they are connected in reverse polarity to the normal rectifier style diode described here, hence their effect on Screen-Grid behaviour and of "sound" is quite different. A real danger with the series Zener Diode configuration is that if the diode breaks down and short-circuits then full supply voltage will be applied directly to the Screen-Grid. This may destroy the tube in the process.

A further (and very effective) enhancement is to use a separate diode for each Screen Grid (or each set in parallel-push-pull) to ensure there is no cross-modulation in the push-pull activity. This places back to back diodes between the Screen Grids, which makes each half of the AC push-pull circuit independent to the other. Less signal averaging and less cross-talk occurs between each half of the push-pull pair, so the sound is cleaner.

In stereophonic amplifiers using a common power supply, this system provides significantly greater channel separation.

Of course, silicon diodes can be retrofitted to an existing amplifier however the negative feedback loop should be re-calibrated to suit the changed output circuit conditions.

It may be necessary to re-calibrate Grid #1 (Control Grid) bias to ensure Plate Current and Plate Dissipation are optimised within the tube manufacturer's design centre ratings.

In already set-up hi-fi amplifiers, it should not be necessary to change operating conditions because the Plate current is already determined by the Screen-grid voltage and not the Plate voltage - but a prudent owner will check in any case to be sure.

If any reader can shed further light on this breakthrough new concept please email your thoughts.

Important: Please note this modification is not suited to Class B guitar amplifier applications (this means most of the "big" guitar amps) because the sound produced by the silicon diode to Screen-Grids configuration is cleaner and less distorted,  dynamic range (transient response) is substantially improved and power output is substantially increased - all advantages for hi-fi but not so good for a guitar amp.

This is because many guitarists, in order to attain or emulate a particular "sound" - and thus for them "normal" usage of their guitar amplifier - operate the amplifier into the severe distortion range by simply driving to full output or more. Some operate in the sustained overload range continuously, using reverb, tape echo, electronic echo or acoustic feedback as a musical effect.

In a typical commercial guitar amplifier - particularly those with tube rectifiers - the power supply will collapse and simply run out of puff when overloaded, resulting in substantially reduced B+ supply voltage, lower power output and increased (severe) distortion as the output signal goes into square-wave like response. But the use of silicon diodes changes the tube characteristics insofar as the normal limit on Plate current as controlled by the Screen-Grid is removed, allowing Plate current to increase in proportion with the signal up to a maximum "saturation" point where more drive in does not produce more power out.

Note also that to maximise output power and minimise distortion in a Class B amplifier, it is vitally essential to balance each half of the push-pull pair of tubes to ensure DC current in the output transformer is reasonably equal. The more tubes in the output stage the harder this is to achieve.

Unfortunately, most high-power (100 W RMS +) guitar amplifiers do not provide individual grid-bias adjustments for the output tubes. Under such conditions, to achieve balanced DC Plate-current it may be necessary to set Grid 1 bias control voltage at maximum signal - not at quiescent (zero signal) as is popularly expounded, and to mix and match the tubes either side of the output transformer to attain reasonably equal balance in total Plate-current per side. (This may result in some unbalanced DC hum at zero signal).

In an amplifier having a single bias supply it may be also necessary to modify the circuit to install a means for balancing AC signal drive voltage into the output tubes because, as well as being dependent upon primary Grid #1 DC bias voltage, the Plate Current will also be dependent upon AC input volts to Grid #1. Note though that the downside to this is that when a tube is replaced, re-biasing is essential.

For example, I conducted a test with a Marshall Model 1959 100W super-lead amplifier, with 4 x EL34 tubes running at 520 VDC B+.  When silicon diodes were fitted to the Screen-Grids instead of the designed 1000 ohm grid-stopper resistor - one to each Screen-Grid, power output increased from about 90W RMS in OEM configuration to about 160 W RMS with diodes. Screen-grid current was negligible, suggesting the tubes behave like triodes, however Plate current increased to about 180 mA per tube, which increased net plate dissipation at full output to about 55 Watts - a certain recipe for very short tube life considering the EL34 has a rated Plate-dissipation of 25 W.

Plate current then in this situation is primarily controlled by Grid #1 alone.

In this case the amplifier simply gets louder and louder without the usual breakdown signs before clipping, enabling the amplifier to be overdriven continuously to self-destruction. In this particular amplifier, the power transformer is rated at about 250 VA, and a quick calculation will show this component will also have a very short life expectancy with silicon-diodes to the Screen-Grids - but the sound is great!!


A conceptual circuit using the Type 803 Power pentode Tube in an OPTIMISED ELECTRON STREAM TECHNOLOGY design as described above, is illustrated below:

This circuit is generic to all 3 Grid power pentodes.

For Tetrodes and Beam Power Tubes just delete the Grid #3 output transformer transformer tap, and its separate Grid #3 DC supply - a Tetrode or Beam Power Tube will have only a Grid 2 in the circuit.

Note: Westinghouse declared the following in their June 1941 user instruction sheet supplied with each 803 tube:

Note the reference to "Tetrode" connection and its effects.

This design feature offers a range of options to the DIY designer/constructor.

Important Note: In the case of a Beam Power Tube, the beam forming plates in a Beam Power Tube are not normally connected to an active circuit element - ie are usually internally connected within the tube to its own Cathode by the tube manufacturer. If available as a separate connection, they should be externally connected to the Cathode in the usual way.

Note: If a voltage measurement is taken either side of the silicon diode in an ultra-linear configuration circuit - ie on the screen-grid side and on the transformer side - obviously a reading will be evident. The reading on one side will be out of phase with the reading on the other side. DC current through the diode can be measured by inserting a small resistor (10 ohms) in series with the diode on the supply side and reading the current through the resistor.





5.1.1    Non-polarised Bypass Capacitor

Irrespective of choice of Screen-Grid and Suppressor-Grid DC supply method, to obtain the full benefit from the application of  OPTIMISED ELECTRON STREAM TECHNOLOGY in Tetrode, Pentode and Beam Power Tube wide-band amplifiers, it is absolutely essential to bypass (or wholly replace) the final polarised electrolytic capacitor in ALL of the B+ Plate AND Screen Grid AND Suppressor Grid supplies, with a suitable non-polarised high quality mica, polyester, polypropylene or oil-filled paper etc. capacitor, having a suitable value (of say 10 to 100 uF for audio), to provide an efficient and stable AC bypass at all signal frequencies and under all operating conditions.

The reason for the use of a non-polarised capacitor is simply that polarised electrolytic capacitors offer asymmetrical impedance to the current flow - depending upon the direction of the current through the capacitor. Most importantly for high-fidelity reproduction, polarised capacitors offer symmetrical characteristics to forward or reverse AC current. This is evidenced by the widespread use of polarised capacitors in loudspeaker crossover networks (which are purely AC) - even in the cheapest commercial speaker systems.

The reason for the relatively large value of capacitor suggested is that it is in series with the load on the Plate and Grids, forming an LC or RC (depending upon the output stage configuration) series network. If the value of C is low then peaks or resonance can occur in the audio range - particularly in the mid to high frequency band where harmonics are present.

This capacitor serves to effectively AC short-circuit (or bypass) the DC power supply and thus eliminate the power supply and its components from the AC signal path, and to ensure that any shortcomings in polarised electrolytic capacitor performance are eliminated - but in such a way that the signal is not significantly aurally affected.

In the case of the Plate circuit of Triode, Tetrode, Pentode, Beam Power Tube and Ultra-linear wide-band amplifiers, the non-polarised capacitor should be installed at the centre-tap of the output transformer.

In the case of amplifiers having separate Plate, Screen-grid and Suppressor-grid DC power supplies this requirement also applies to each of the Screen-grid and Suppressor-grid DC circuits as applicable.

To ensure adequate low frequency response each of the separate circuits must have as large a value as is practicable.

A 400 VAC (560 VDCW) Motor Run Capacitor is ideal for this function and they are readily available in non-polarised polypropylene construction. The voltage rating must be adequate to handle the sum of the B+ DC voltage plus the AC rms signal output voltage.

I would recommend a parallel connected non-polarised capacitor arrangement, to provide 100 to 200 uF total. For higher B+ voltages the capacitors can be connected in series (800 VAC 1120 VDCW), noting that when seriesed, the effective capacitance is halved.

A further refinement is to apply the "Rule of Hundredth's", which says that instead of using a single capacitor - eg a large electrolytic, a bank of capacitors is installed. Each capacitor is one hundredth the value of its adjoining capacitor. For example, instead of installing one single 100 uF electrolytic, instal a 100 uF, 1 uF, 0.01 uF, 0.0001 uF etc. wired in parallel. Lead length of the smaller capacitors must be kept a short as possible to prevent stray capacitance or inductance. It is also important to ensure all capacitors installed are capable of coping with the applied voltages without stress.

Of course the bypass capacitor is installed on the line (source) side of the Screen-Grid-stopper resistor and/or diode, as the case may be.

Note: This method may result in resonances in the audio range, particularly to harmonics, resulting in sibilant accentuation and "hissy" voice. It is thus not suitable in all amplifiers.

Note 1: Please note that the irrespective of the value of the electrolytic filter caps - even to many thousands of uF - the value of the non-polarised cap is still critical in relation to tone, or spectral balance, across the audio range. To optimise tone experimentation is essential.
Note 2: Modern "fast" capacitors have a different tone to older oil-filled types. It may be necessary to use oil-filled paper caps to obtain a smoother, less harsh tone. It all depends upon the circuit design and componentry used. There is no definitive answer. Unfortunately the older oil-filled paper caps are physically larger so need more chassis space. They also do not usually come with flying leads, which makes wiring more difficult because the terminal lugs are exposed. Ensure voltage rating is adequate.

5.1.2    Electrolytic Bypass Capacitors to B+ Supply

Please note that as explained in my Power Supplies page, notwithstanding the obvious benefits from the use of non-polarised capacitors as described above, it remains essential to good transient response to instal large values of electrolytic capacitors to store adequate power to satisfy the demands for transient peak signals.

The question is "how big a value"?

In the case of power requirements, full particulars are provided in my Power Supplies page.

But what about "sound"?

It has been demonstrated by early amplifier designs produced in the 1940's and 1950's that the frequency response - even in the highest quality equipment, tended to roll-off at both high and low frequencies.

Frequency response usually deteriorated as power output increased. This was an attribute ("power response") not usually presented in the glossy sales brochures.

Modern digital recording techniques and playback media have dramatically increased both low and high-frequency response for the typical recording, challenging even the very best of tube audio amplifier designs - particularly when played at high-volume.

But all is not lost!!

Modern capacitors offer improved performance and reliability over their ancestors. They are also available in values unheard of in the days of tube rectifiers. (eg 32 uF was "large" in the 1950's but now 100,000 uF is "medium.")

The thin film caps of today charge and discharge much faster, enabling larger values to be used in common applications where large values were previously forbidden - such as interstage coupling caps.

Modern capacitors offer reduced unwanted side effects, such as inductance, leakage and resonance, so offer improved high-frequency performance and audio clarity.

One effect of this is a reduction in the level of negative loop feedback needed to offset roll-off in frequency response of an amplifier - a definite advantage.

In the case of low frequency performance we can retrospectively improve old designs by using higher values of filter bypass capacitors.

Since the bypass capacitor forms the return AC circuit for each stage in the amplifier, it follows that the impedance of the bypass (filter) capacitor at any given frequency will be a portion of the total impedance of the circuit being bypassed. Now since the impedance of a capacitor varies in direct proportion with the value of capacitance at any given frequency, it follows that providing we reduce the value of the capacitor's impedance to a value that has minimal effect upon the circuit, then we can attain improved performance from the circuit.

Another way of explaining the concept is that traditional tube electronic design engineering principles assume that the B+ rail is at AZ zero (or earth) potential - regardless of the applied DC voltage.

This is a practical approach for many applications, but ignores the reality that capacitors - particularly those of the electrolytic variety - have their own characteristics, which are injected into the circuit and therefore MUST influence the sound we hear.

Since the capacitive reactance of a 25 uF capacitor at 30 Hz is nominally only 212 Ohms, it suggests that the influence of the capacitor may be negligible. However if the inductance of the capacitor is only 0.1H then the inductive reactance in the capacitor is 3140 Ohms at 5 kHz - and more at higher frequencies.

So we can see that increasing capacitive reactance will decrease low frequency performance and decreasing capacitor induced inductive reactance will increase high-frequency performance.

Since measured inductance and capacitance values from typical large capacitors - eg above 500 uF - do not appear to display the characteristics experienced aurally there may be an alternative explanation. It may be that the bypass capacitor forms an RC or LC series network with the plate resistor and/or output transformer, forming a bass boost circuit. This is easier to see with the output transformer, where either one end (in the case of a single-ended output stage) or the centre-tap (in a push-pull output stage) forms the terminal at which the output is taken off. but in the case of driver stage it would seem the series decoupling resistor causes the formation of the RC network that creates the bass boost effect.

Either way it is desirable to tune the network to the preferred frequency - eg 40 Hz

The question is what is the magic number?? How big to we need to go??

To attain extremely good low-frequency response, from practical experimentation I would suggest the minimum value of the bypass capacitor for any stage in an amplifier or pre-amplifier, including the output power stage, may be calculated by dividing the constant 15,000,000 (15 million) by the value of the plate load resistor (or cathode-follower cathode load resistor) for that stage.

This approach will produce values of 150 uF for a 100k plate resistor and 2,250 uF for a 6,600 Ohms Plate to Plate output stage.

If those values appear frightening, then try a lesser constant - say 10,000,000

If the B+ supply supplies more than one stage or, in the case of some phase-splitters having more than one plate resistor, then the value of the resistor should be calculated as the average of all the resistors in the circuit.

If the B+ supply supplies a stereo pair of amplifiers then the value of the plate load resistor used to calculate should be half of the single channel value - ie each bypass capacitor should be twice the size as for a single channel.

If the output stage uses more than one pair of tubes from a single transformer, then use the transformer actual nominal rated load impedance - not the effective impedance as seen by each pair.

It will be noticed that these values of capacitance are substantially higher than convention, however this is what I have determined form extensive critical listening tests.

The general object of this design approach is to introduce a circuit resonance induced tonal characteristic that is pleasing.

Too high a value of capacitor will deliver undefined bass, so some tweaking may be necessary.

It is desirable for all stages to have the same tonal characteristic, so this formula assists to achieve that. In other words, each stage should ideally have proportionately the same tonal or frequency characteristic.

Note 1:  This "rule of thumb" formula deals only with tonal characteristics - power (energy) requirements for the output stage are not covered by this. However it is easily seen that the values produced by this design approach will supply adequate power for most applications.

Note 2: Regardless of the value of electrolytic capacitors used for bypassing and filtering, non-polarised bypass capacitors are still essential for good high-frequency performance and low intermodulation distortion.

Note 3: In the case of guitar amplifiers the above formula is not applicable because the lowest frequency to be reproduced is about 80 Hz. In this case a constant of say 10,000,000 or less would be appropriate.

Note 4: In the case of bass guitar amplifiers, where the lowest frequency to be reproduced is around 40 Hz, then for outstanding results the constant needs to be in the region of 20,000,000. This value of capacitance will also deliver adequate power to the output stage.

Note 4: Large values of capacitor can be lethal, so discharge resistors should be installed as per instructions provided in my Power Supplies page.

Note 5: In my experience, larger bypass capacitors have the effect of improving circuit stability and improving high-frequency audio performance.



5.2.1     Introduction

Since the advent of Tetrodes and Pentodes, it has been standard practice to instal a small bypass capacitor from the Screen-grid to ground (Cathode) in Tetrodes and Pentodes used in voltage amplifier stages.

This is shown as C2 in the figure below.

The purpose of the bypass capacitor is to reduce Grid to Plate capacitance, remove undesirable audio and high-frequency signals such as RF components from the output before the load, and to improve the decoupling and stability of the stage.

In voltage amplifier (driver) stages the value of this capacitor has historically been in the order of 0.1 to 0.5 uF, the latter value being considered by Radio Engineers to be adequate for good quality audio purposes. A similar situation exists in RF power amplifier stages.

It should be noted though, that this value of capacitor is usually associated with small tubes in high-impedance circuits, such as the EF86, 6AU6, 6U8, 7199 etc, where the value of Screen-grid supply resistor may be in the range 100k to 1 Meg Ohms with very low Screen-grid current, .

An example is shown in the Mullard High-fidelity Pre-amplifier.

5.2.2    Theory

For the theoretically minded, the formula for calculating the value of the bypass capacitor in a voltage amplifier stage is given courtesy of the Radiotron Designers Handbook, 3 rd Edition (1940).

5.2.3    High-fidelity Audio POWER AMPLIFIER Applications

Notwithstanding the above historic convention, in high-fidelity audio POWER AMPLIFIER output stages a very different situation applies.

If we think for a moment, it can be easily seen that there is both a DC and an AC signal path from the negative terminal of the bypass capacitor up through the capacitor to the Screen-grid external to the tube, thence from the Screen-grid to the Plate inside the tube. The latter will be the case (even if we do not want it) because the Screen-grid is negative to the Plate.

As explained above, the portion of Screen-grid current in a power tube can be quite high - depending upon output stage configuration and applied voltages to the Screen-grid and Plate respectively.

Since the internal impedance of the bypass capacitor will be relatively small and the Screen-grid is operating independently of Control Grid #1, it follows that the magnitude of the current flowing from Screen-grid to Plate will depend more or less entirely upon the actual applied DC voltage between Screen-grid and Plate and the value of the Plate load - which will be also seen by this secondary circuit.

That is "secondary" to the primary Cathode to Plate circuit.

Obviously, the higher the difference between actual applied DC voltage between Screen-grid and Plate, the more current will flow.

In Tetrode and Pentode and Beam Power Tube applications where the Plate and Screen-grid operate at the same DC voltage, including Triode and Ultra-linear connections, in the conventional and very common configuration shown above, it is suggested by most writers that the Plate will function as the primary anode so long as the Plate signal voltage does not drop below the DC Screen-grid voltage. In this case, tube manuals show that about 10-20% of Cathode current is lost in the Screen-grid circuit.

Since the bypass capacitor C1 is common to both Plate and screen-grid circuits, in terms of frequency response and dynamic response whatever happens in one will happen in the other.

But when the DC Screen-grid voltage is less than the Plate voltage - as in RF linear amplifiers, transmitter modulator amplifiers, transmitting tube audio amplifiers, Public Address amplifiers or OPTIMISED ELECTRON STREAM TECHNOLOGY applications as shown in the following diagram - then current MUST flow between Screen-grid and Plate.

This is because a secondary DC and AC circuit is established between Screen-grid and Plate, with the Screen-grid forming the negative terminal/element.

In high-voltage amplifier designs, the voltage between Screen-grid and Plate may be in the order of several kilovolts (kV).

Since the lower leg from Cathode to Screen-grid is through the bypass capacitor, and that is external to the tube, it follows that the limiting factors to current flow in this secondary circuit will be the Plate load impedance for AC current (the DC resistance of the output transformer is negligible) and the value of the Screen-grid resistor if used (either voltage dropping resistor or grid-stopper) for both AC and DC current.

For explanatory purposes, the internal tube resistance between Screen-grid and Plate may be assumed to be zero.

It follows then that the Cathode to Screen to Plate secondary AC circuit is in parallel with the Cathode to Plate primary AC circuit.

As explained above, the value of Screen-grid current - and therefore its contribution to audio power output, can be significant.

Since the current in both circuits combine together in the negative to positive Screen-grid to Plate section of the circuit inside the tube (and thence common return to the Cathode via the B+ filter capacitor at the output transformer centre-tap) it follows that any difference in the audio signal between primary and secondary circuits will be apparent at the output transformer.

For example, if the Screen-grid bypass capacitor is too small, that portion of audio power output contributed to by the Screen-grid secondary circuit as described herein, will not have the same low-frequency response as the primary Plate circuit and therefore low-frequency power output will be proportionately reduced.

For example, in the Mullard High-fidelity Pre-amplifier circuit, typical of conventional design, the Screen-grid bypass capacitors C9 is 80 times the value of C8, and C17 is 160 times the value of C12.

The value of the Screen-grid bypass capacitor will also affect the operation of the Plate circuit B+ bypass capacitor, because the B+ Screen bypass cap and B+ Plate bypass cap are in series in the signal circuit.

Hence it may be deduced that:

Where a separate Screen-grid power supply is provided,  it is most important for full-power hi-fi reproduction at very low frequencies, and for signal balance within the tube, to ensure the value of the final Screen-grid B+ bypass filter cap is not less than the value of the Plate circuit B+ bypass filter cap.

In the above diagram the Plate bypass capacitor is shown as C1 and the Screen-grid bypass Capacitor as C2.

Note: Even though the installation of a voltage dropping resistor or grid-stopper resistor to the Screen-grid (R3 in the voltage amplifier circuit diagram above) may reduce AC and DC current, the issue of AC impedance remains - hence the grid resistor may be disregarded for this aspect of hi-fi design.

It will be readily seen that in a power output stage where the Plate and Screen-grid share a common AC circuit return at the power supply, and regardless of the use of a grid-stopper or dropping resistor (R3 in the above diagram) or not, the above argument does not apply because both Plate and Screen-grid will share a common B+ bypass capacitor.

BUT - where the screen-grid is supplied through a filter choke from the centre-tap of the output transformer at C1, and bypassed by its own electrolytic capacitor C2, the configuration shown above is still is applicable.

However bear in mind that the regulation of Screen-grid voltage is also extremely important so a separate supply should remain an essential design element.

The above comments also apply to the Suppressor-grid

5.2.4    Cathode Bypass Capacitor

In the case of Tetrode or Pentode amplifier output stages having Cathode Bias, it is usual to instal a Cathode-bypass Capacitor as shown in the following diagramme:

It can be readily seen from the way the circuit is drawn that the Cathode-bypass Capacitor is in series with the B+ Plate and Screen-grid supply.

It follows that if the Cathode-bypass capacitor is too small, that portion of audio frequency response determined by the Cathode circuit will not have the same low-frequency response as the primary Plate circuit and therefore low-frequency power output will be proportionately reduced.

From the above explanations it can be demonstrated that the value of C3 must be at least equal to the value of C1.

The more theoretically minded can calculate the actual values needed for equilibrium in the frequency response characteristics for each part of the circuit.

Where a separate Screen-grid power supply is used, as shown below,

it can be readily seen that the Plate and Screen-grid bypass capacitors C1 and C2 are in AC parallel but the set of both is in series with the cathode bypass capacitor C3.

Hence it is also essential that the nominal capacitance value of C3 is equal to the SUM of C1 + C2.

Where separate Cathode-bypass capacitors are used to each Cathode, the above rule still applies for each capacitor.

It is usual for Class A amplifiers to use a single common cathode resistor and no bypass capacitor per push-pull pair of tubes - this is obviously more AC linear.

5.2.5 Power Factor

Ideal values for C1 and C2 and C3 can be in the region of up to 5,000 uF per push-pull pair of  tubes - see warning re risk of electrocution.

To maintain OPTIMISED ELECTRON STREAM TECHNOLOGY design standards in the output stage circuit, some circuit tuning is essential to optimise the "POWER FACTOR" in the output stage circuit.

In terms of "sound", this is a very critical issue.

This extra step is needed to tune resonances in the output stage circuit to match the characteristics of the output tubes, output transformer and loudspeakers to the signal "sound" or "tone" and listening room acoustics. Too low a resonance may result in a "dull" or "flat" bass sound. Too high a resonance may result in "bass boom" and loss of definition.

To optimise the output stage circuit and deliver maximum power from minimum output impedance with maximum loudspeaker damping characteristics, changes in power factor produced by the inductive reactance in the output transformer will ideally be cancelled out by the capacitive reactance in C1.

To optimise the value of C1 some degree of practical "trial and error" experimentation is needed - a frustrating experience but one worth the effort.

Note C2 and C3 need to be suitably modified as described above and hereunder to maintain the correct ratio to C1 - ie the same capacitance value.

Note also that to overcome non-linearity in the AC bypass circuit, C1 and C2 may need to be bypassed by a small stabilising capacitor in the region of 0.5 uF - but since this small capacitor will directly affect the high-frequency response, it too must be chosen with care regarding both size and material of construction.


All of the Plate Circuit output stage power passes through the AC circuit formed by the output transformer and C1 - and whatever else shunts C1.

Since L1 is directly connected to the Power Supply, it follows that the Power Supply is in series with L1 and shunts C1.

This is illustrated in the diagram below:

So to minimise the effects of the Power Supply rectification and filter circuitry on "sound", the value of L1 should be as large as can be practicably sustained - noting that the DC resistance of L1 will directly reduce the available B+ voltage to the Plates.

The larger L1 is, the more AC current will be forced through C1 and less through L1 and associated circuitry.

Another way of expressing this is to say that whatever AC power is lost into L1 and its associated source bypass components - all of which shunt C1 - the less linear the output stage will be at low frequencies.

An inductance of about 10 Henries (minimum) is desirable for L1.

This also holds true for simple one stage filter choke systems, because the rectifier/filter is always shunting C1 - thus will always affect its performance and effect on "sound".

IMPORTANT: To prevent instability in the power supply (which dramatically affects audio sound as heard through the loudspeaker) it is desirable to ensure C2/L1 and C3/L2 are of equal value. Where only C2/L1 are used then it is desirable that C1 and C2 are of equal value. It is thus obvious that a three stage filter enables a higher value of C1 than for a two stage filter.

This point will be expanded in future revisions of this page.

5.2.6    Silicon Diode Feed

When silicon diodes are installed in the Screen-grid and Suppressor-grid circuit when using my OPTIMISED ELECTRON STREAM TECHNOLOGY design, the above bypass requirements still apply because although the diode will block AC from passing from the Cathode to the Screen-grid and Suppressor-grid internally, it cannot stop the current flow in the external secondary circuit described herein.

5.2.7    Voltage Doubler Power Supply

Where the Screen-grid B+ supply is taken from the output of a voltage doubler rectifier, it may be the case that for convenience the Screen-grid supply is taken from the mid-point of the two series-connected filter capacitors.

However if we analyse the effect of this arrangement having regard to the above, it is easy to see that the external Screen-grid bypass current is passing from negative to positive through the lower filter capacitor.

It is also easy to see that return external Screen-grid bypass current is passing from positive to negative through the lower filter capacitor.

Consequently, it is evident that the two currents will cancel out.

Bad move!!

So in other words, this configuration is not desirable for high-fidelity reproduction, even though it may appear to work satisfactorily in public address amplifiers.

There is also an issue of the effect of differences in capacitor characteristics in the forward and reverse directions, as well as the effects of distortion caused by the output transformer.

There is also the issue of excessive ripple in the DC supply.

One solution to both issues may be to instal a 60 mA filter choke in the line (supply) side of the Screen-grid B+, then add a second filter capacitor to the Screen-grid B+ at the load side, to provide a separate path for forward and reverse currents.

The value of the second (final) bypass capacitor should be equal to or greater than the value of the seriesed pair.

5.2.8    Capacitive Voltage Dividers

Where the Screen-grid B+ supply is taken from the centre-point of a pair of series-connected electrolytic capacitors in the Plate B+ circuit, the same problems arise as explained above.

NOTE: To ensure equal voltage distribution over time (service life) series connected electrolytics MUST be stabilised for equal voltage distribution by means of an external circuit system such as shunt resistors (voltage divider network) or power supply configuration.


In the case of OPTIMISED ELECTRON STREAM TECHNOLOGY where a separate Screen-grid and Suppressor-grid power supply is provided, for full-power hi-fi reproduction at very low frequencies it is essential to ensure the value of the final Screen-grid B+ bypass filter cap (and Suppressor-grid in the case of a Pentode) is NOT LESS THAN the total effective value of the the sum of all Plate circuit B+ bypass filter capacitors directly connected to the centre-tap of the output transformer - polarised and/or non-polarised.

Where Cathode-bias is employed, in addition to the above it is also essential to ensure the value of the Cathode-bypass capacitor(s) is not less than the value of the Plate and screen-grid supply capacitors

Note: Where one or more filter chokes are used in the B+ supply, the value of capacitors before the choke(s) may be ignored for this requirement.


Where negative loop feedback is used from the loudspeaker terminals back to an early voltage amplifying stage, it can be seen from the above that any defects in performance in the output stage will simply be transferred back to the input, with the result that the amplifier will always be in a constant state of correcting itself.

In real life we have the following variables in the power stage system:

Hence any use of negative loop feedback should be deferred until the output stage is optimised according to the requirements of my OPTIMISED ELECTRON STREAM TECHNOLOGY

The best plan is to forget it and live with the best option without it.

Please refer to my SCREEN-GRIDS page for full particulars of this requirement.




In output stages of audio amplifiers it is common to use FIXED BIAS or BACK BIAS to apply and control the Grid #1 - Control Grid bias voltage in the output tubes.

Designers always consider the requirements for direct current operating conditions but often ignore requirements for alternating current conditions - ie signal voltage.

The Grid #1 Resistor to each output power tube forms part of the load for the preceding driver stage. Hence for maximum efficiency and stability, the return path from the output of the driver stage circuit back to its Cathodes should be direct and have very low impedance.


Bias, by definition, requires a voltage potential to be present between the Control Grid and the Cathode.

CATHODE BIAS usually results in the Control Grid being at nominally 0 VDC and the Cathode at the required bias voltage being + VDC.

FIXED BIAS usually results in the Control Grid being set at the required bias voltage being - VDC and the Cathode at nominally 0 VDC. In the case of FIXED BIAS it is essential to bridge the difference between the central axis of the AC signal input and the power tube Cathodes, such that the central axis is at the same AC voltage as the power tube Cathodes.

It is essential for stable high fidelity operation to ensure that:

a)    the Grid #1 circuit has a reliable and predictable low-impedance return path for the AC signal voltage from the preceding stage
b)    the Grid #1 circuit has a reliable and predictable low-resistance return path for the DC bias voltage between Grid #1 and the Cathode of the same tube.
c)     in a balanced driver stage for Classes AB, AB1, AB2 and B, where the power tubes are biased towards cut-off and driver stage output signal voltages are symmetrically balanced about a virtual central axis having a nominal potential of 0 VAC, it is imperative to ensure the junction of the two Control Grid resistors of the output tubes be AC grounded to 0 VAC.

Thus in an RC (resistor/capacitor) coupled amplifier, it is a fundamental requirement that the Grid #1 Resistor to each power tube provides a return circuit path to the Cathode of  BOTH the preceding driver tube AND the power tube to which it is connected.

This is usually, but not always, through the earth or ground terminal of both AC and DC applied circuits.

Fortunately, in most amplifiers using FIXED BIAS or BACK BIAS, the Cathodes of the output tubes are directly earthed (to the chassis), thereby providing a convenient return circuit at 0 VAC potential.

6.2    Cathode Bias

In most amplifiers using CATHODE BIAS, the Cathodes of the driver tubes are earthed either through a bypassed or unbypassed Cathode resistor, hence the Cathode terminal is indirectly AC earthed to complete the return circuit for the driver tube(s).

This principle applies particularly to CATHODE BIASING of power tubes, where the Cathode terminals may well be at say +50 VDC but simultaneously at  0 VAC. . (An exception is where the output stage is in Class A and the common cathode resistor is unbypassed to develop negative current feedback in the Cathode circuit.)

Typical Ultra-linear output stage configuration with CATHODE BIAS. Note the bypass capacitor on the common Cathode resistor, thereby providing a low-impedance AC path to ground.

Typical Triode-connected Pentode output stage configuration with adjustable CATHODE BIAS.
Note the absence of a bypass capacitor on the common Cathode resistor.

This circuit features + or - adjustment of common bias voltage and also fine adjustment of balance between the tubes.

6.3    Fixed Bias Adjustment

It is common practice in FIXED BIAS amplifiers to incorporate an adjustable bias control circuit, incorporating an adjustable resistive network to enable precision adjustment to the Control Grid voltage and/or Plate Current of the output power tubes.

Typically, this negative polarity DC voltage is sourced from a half-wave or full-wave tube or solid-state rectifier, filtered by a simple resistor/capacitor network.

Typical Fixed Bias amplifier incorporating adjustable bias to the power tubes.

Note: In this design, the 20 MF (uF) capacitor serves to bypass the AC return circuit from the driver stage to ground, thereby providing a direct return circuit path to the Cathodes of the driver tube. This capacitor is connected in such a way as to ensure the Control Grids are equally earthed at all times through their respective Grid Resistors, regardless of applied DC bias voltage. It also ensures the central axis of the balanced signal input is at the same potential as the Cathodes - an extremely important component.

6.4    Bypass Capacitor Material

In commercial amplifiers with FIXED BIAS, to minimise cost the bypass capacitor is nearly always of the polarised electrolytic variety, but this means that the AC return circuit is not symmetrical. (Electrolytic capacitors have different characteristics in positive and negative polarity circuits).

Consequently, to ensure symmetrical AC circuit configuration it is essential to instal a suitable non-polarised bypass capacitor into the circuit at a point closest to the Grid #1 of the output tubes as is practicable.

Typically this will be at the junction of the two (or more) Grid Resistors.

If a driver transformer is used, and its centre-tapped secondary is not directly connected to the power tube Cathode circuit, then instal the bypass capacitor between the centre tap of the secondary and earth.

It is therefore absolutely essential to bypass (or wholly replace) the final polarised electrolytic capacitor, with a suitable non-polarised high quality mica, polyester, polypropylene, paper or oil-filled paper capacitor, having a suitable value (of say 1.0 uF or a 10 uF or more motor start capacitor as above for audio), to provide an AC bypass at all signal frequencies and under all operating conditions.

This small capacitor serves to effectively short-circuit (shunt or bypass) the DC power supply and thus eliminate the power supply and its components from the AC signal path of both the driver tube and power output tubes, to ensure that any shortcomings in polarised electrolytic capacitor performance are compensated - but in such a way that the signal is not significantly aurally affected.

Importantly, it also provides an automatic safeguard against the adverse effects of poorly contacting bias potentiometers and/or adjustable wire-wound resistors.

Normally, the value of the bias supply capacitor will not exceed 10 uF (to ensure fast charging to full bias voltage before the output tubes heat up and commence to conduct Cathode Current) so an extra one or two uF will not significantly affect the charging circuit performance.

If preferred, to achieve the same end result as described above, the final Control-grid power supply filter polarised electrolytic capacitor (and the first as well if so inclined) can be wholly replaced with a polyester or paper or motor start capacitor of say 8 uF value and having a suitable DC voltage rating.

Suitable capacitors may also be reclaimed from unwanted fluorescent lamp-holder assemblies.

For the ultra-fastidious, the "Rule of Hundredth's" may also be applied.

Note: There is no practical limit as to the value of the bypass capacitor, provided the bias supply is capable of charging it quickly to ensure bias voltage is present when the power tubes warm up and commence conducting. I have successfully used values around 100,000 uF, shunted by suitable non-polarised polypropylene caps.




The concept is shown here:

The purpose of this configuration is to stabilise the voltage into the B+ rail. (Note: "Stabilise" means free from self-oscillation)

Ripple voltage appears at the input.

The ripple voltage includes harmonics from the mains and spikes from the rectifiers - both of which pass through the inductor filter choke and appear at the output transformer centre-tap (AC earth) - so thereby appear in the output stage circuit as a signal modulating voltage and, by transformation, audible in the loudspeaker as distortion.

The shunt diode short-circuits the choke in the reverse direction

Any unwanted spurious voltage - such as switching spikes - are either eliminated from the system or dramatically attenuated

The use of a small capacitor or capacitors connected in parallel with the first electrolytic cap provides a path for spikes and any other mains harmonics etc to be diverted to the negative rail.

The output voltage is therefore, for practical purposes, spike free

The capacitors should be of equal value and type so that the two capacitors operate in sync with the input voltage - ie the current in is matched by an equivalent current out, synchronised in time. That is to say the charge and discharge rates over time should be equal. If they are not equal, higher than normal rectified ripple mains frequency (ie 100 Hz or 120 Hz) ripple spikes can be generated within the filter system.

Capacitance values up to 20,000 uF work well for me.

NOTE: See warnings re risk of FATAL electric shock from charged capacitors.

The benefit of the PI filter is that even a small value of choke inductance does wonders for substantially reducing ripple.

Elimination of Ripple is vitally important for true hi-fi because the B+ rail is the AC ground for the signal.

If this voltage "dances" around with spurious ripple signals and/or the actual centre-point of the output transformer not being at its nominal zero AC volts true centre, then the output will be modulated by this voltage. It can either be bypassed to ground via the filter capacitors or transformed to the loudspeaker (in which case it will appear in the negative feedback system - if used)

This system is very stable and enhances clarity and frequency response




The conventional B+ supply looks something like the following diagram:

It is explained above why Filter Capacitor C1 forms a vital part of the AC circuit in the output stage.

It is also explained why it is desirable to "optimise" the value of C1.

Further, it is explained that the Power Supply is connected in parallel with C1 and therefore also forms part of the AC circuit of the output stage.

Consequently, it can be easily demonstrated that the "sound" of the amplifier will be directly affected - maybe adversely - by the characteristics of the Power Supply.

Since the object of OPTIMISED ELECTRON STREAM TECHNOLOGY is to "optimise" both the performance and sound of the amplifier, it follows that if the Power Supply can be isolated, or decoupled, from the power stage then a positive benefit might result.

This object may be easily and economically realised by installing a series connected silicon diode rectifier into the B+ supply - having its arrow pointing towards the load - between the Power Supply output terminals and C1 - ie AFTER  the Power Supply and BEFORE C1.

Note that C1 must always remain part of the Amplifier, because it forms a direct path in the AC output circuit.

So regardless of whether the Power Supply is of the simple centre-tapped full-wave or full-wave bridge rectifier or full-wave voltage doubler rectifier, or it has a capacitor filter, choke filter or some other combination of filter - or even includes a voltage stabiliser or regulator - to ensure the OPTIMISED VALUE OF C1 is not altered by shunting the Power Supply circuit, significant improvement to performance can be made by installing a series silicon rectifier diode as described.

To ensure cool operation and reliability, the current rating of the rectifier should be generous in relation to the power requirements of the amplifier - eg 3A or 6A. The rectifier can be of the same specification as that used in the Power Supply B+.

Where a separate Screen-grid Power Supply is used, then the same principles apply to C2.

Where an inductor - such as L1 above - is used in the Power Supply, it is preferable for the diode to be installed after the inductor - to ensure the Power Supply is completely isolated from the amplifier.

This method makes it easier to adjust the tonal balance or "tone" of the amplifier power stage - because fewer components are now in the signal path - therefore there are fewer interactive variables to deal with when changing component values.

The same principle apply to L1 and C2.

It will be seen for the above that in the case of simple filters having just one capacitor after the rectifier - and nothing else - the requirement described is met. Unfortunately that simple configuration results in significant hum and ripple and poor quality direct current (DC) - so is not recommended for high-fidelity amplifiers.



Having described how decoupling of the amplifier from its power source facilitates and enhances performance, it follows that the same principles apply to driver stages.

In a conventional amplifier, the stages are connected in a "cascade" configuration.

That is to say, each stage "cascades" into the next - just as a stream "cascades" over its riffles as it runs down a slope.

Conventional theory says that up to three stages may be connected to the same B+ supply before instability becomes a problem.


For optimum performance each stage should be a self-contained discrete circuit - wholly independent from those before and those after.

The usual method of decoupling is to instal a series resistor into the B+ supply between stages. But this produces significant voltage drop when all or any tubes draw current over the steady state condition.

This convention also allows signal from one stage being conducted to an adjoining stage when there is a signal voltage difference between those stages.

So to produce effective decoupling between stages and to prevent each stage sharing the bypass capacitors and associated circuitry installed in adjoining stages - before or after - it is essential to decouple effectively and completely.

To understand the OPTIMISED ELECTRON STREAM TECHNOLOGY: SERIES DIODE DECOUPLING concept first we need to consider the amplifier as a "system"

A "system" is defines as a series of "processes", each having an input and an output. The ideal system is self-adjusting or
self-correcting, by means of a feedback loop (s).

It follows that in such a system what happens in one part also happens in others.

In the case of an amplifier, the signal is present on the B+ rail at each individual stage. In some cases the signal is in phase with other stages, and in other cases the signal is out of phase with other stages.

Therefore the B+ rail is common to all stages. Looking at it another way (ie from the B+ line), the B+ rail is the output from a simple
parallel mixer system - each stage being an input to the mixer.

Clearly then the number of stages and those in phase and those out of phase will affect the "sound" of the amplifier.

Conventional design principles regard the junction between the plate load resistor and the immediately adjacent filter cap as being at AC "earth" - because the filter cap is regarded as being a very low (insignificant) impedance path for the AC in the circuit

However practical tests reveal a different story

I submit any given electrolytic cap has different characteristics when comparing its forward and reverse current flow. One way to
demonstrate this is to replace an electrolytic in the B+ rail with a non-polarised motor start cap - the difference in tonal quality is

This proves that the junction of plate resistor and filter (signal bypass) cap is not at AC earth at all, but at some point above it - or at best the AC signal circuit will be affected or influenced by the series connected bypass capacitor's inherent internal characteristics.

Now since we are talking a VOLTAGE amplifier we do not need to consider the CURRENT in the B+ rail but the VOLTAGE


When a transient signal appears on the B+ rail at any point it will appear across the entire rail - not just at the stage in which it

So if, for example, the phase-splitter B+ voltage momentarily sags, the signal from the previous stage - which does not suffer as
much sag because it is a low current stage - will flow to the phase-splitter stage and be mixed with it at the filter cap

Ohm's Law applies, so wherever there is a voltage difference you will see current flow - in whatever direction it chooses

The purpose of the series diode in the B+ rail is to prevent SIGNAL voltage from transferring forwards (positive feedback)  in the
circuit whenever a voltage difference in the B+ rail appears

It will not of course reduce negative feedback between stages through the B+ rail

Provided a suitable diode is installed to each and every stage in this way, this simple device prevents the AC in any single-stage circuit from being shared by an adjoining upstream (line-side) stage - thus ensuring all of the signal current is passed through the stage discrete bypass (usually "electrolytic") filter capacitor.

Thus each stage can be optimised in its own right.


This element of OPTIMISED ELECTRON STREAM TECHNOLOGY describes the design principle wherein the AC signal drive system circuit comprises a symmetrical balanced AC signal drive system whereby the AC signal OUTPUT from the Driving stage circuit and the AC signal INPUT of the Driven stage circuit must both be directly connected to a common central axis at reference voltage potential.

That is to say, in a push-pull amplifier stage, the mid-point central axis of the AC push-pull signal input voltage MUST be directly connected to the mid-point central axis of the the AC push-pull signal output voltage - ie the INPUT voltage to the driven stage.

Transformer Coupling/Driver System

In a conventional transformer coupled circuit, the supply terminal of the single-ended driving side, or centre-tap of the push-pull driving side is earthed via the B+ bypass cap.

In this case, the return circuit for the AC signal is through the transformer primary back to the driver cathode via that B+ bypass cap.

Being connected to earth it follows that the driver cathode bias resistor should be also connected to earth. In the diagram below that would be via a centre-tap on the driver Triode AC filament transformer.

In some Class B designs using zero bias tubes, and in a cathode bias design, the centre-tap is usually connected to ground as shown.

But in a fixed bias design, the centre-tap of the transformer secondary is usually connected to the bias supply.

Thus when the centre tap of the secondary is connected to the bias supply then the primary and secondary centre-taps do not share a common connection - except through the bias supply earth point, which is at positive polarity in respect to the bias voltage.

This can be seen in the following design, where the single-ended driver is coupled to a push-pull output stage.

Note the cathode bias resistor and bypass capacitor in the driver stage are grounded but the centre-tap of the transformer
secondary is connected to the negative voltage DC bias supply.

But being a transformer none of that matters, because the primary and secondary are two completely independent and
isolated circuits.

RC Coupled Driver Stage

However when a tube RC coupled driver circuit is used with either plate or cathode output - then the two circuits must share a common in and out AC signal axis - or else the sound will be affected.

This is explained below.

Fig 1: Cathode Biased Output Stage

In a cathode biased amplifier the requirement for driving and driven circuits to share a common connection is normal, conventional design practice. A typical standard configuration is shown in Fig 1.

Notice how the central axis of the driving and driven circuits is common. In this case the central axis is at ground potential.

Since the circuitry in both halves of the push-pull circuit are symmetrical and exactly equal it follows that the AC signal voltages in both halves of the push-pull circuit are symmetrical and exactly equal.

However this is not the case with fixed bias amplifiers.

Fig 2 and 3: Fixed Biased Output Stage

Fig 2 shows a conventional fixed bias circuit.

Fig 3 shows a conventional fixed bias circuit with the bias supply shown.

Notice how these circuits also appear to be balanced each side of the push-pull central axis.

However these designs are not balanced and are not symmetrical.

Further explanation is provided below.

Fig 4: OPTIMISED ELECTRON STREAM TECHNOLOGY Fixed Bias Symmetrical Balanced Drive System.

Notice how this circuit appears to be the same as Fig's 2 and 3.

However there is a subtle difference.

Notice how the Cathode circuit of the driving stage is not connected to ground - as is conventional design practice - but to the bias supply for the driven stage.

This is the profound difference.

Further explanation is provided below.

Fig 6: Conventional Fixed Bias System.

Fig 6 shows the AC signal path in a conventional fixed bias system.

Notice how the AC signal must pass through the bias supply components - particularly the bypass electrolytic filter capacitor (usually of the electrolytic variety). In this case, the bias supply shown is simple, but in most designs the bias supply is complex and often comprises chokes and other harmonic producing or refining components - in complex inter-relationships.

Moreover, it is common practice to use half-wave rectification in bias supplies, so the residual ripple voltage will appear as a series of pulses in series with the grid driving circuit.

Consideration must also be given to the effects of spurious harmonic and transient spike signals injected into the bias circuit from the bias supply mains transformer.

Since this part of the driving circuit is common to both halves of the push-pull drive, it follows that the signal will ALWAYS be modified by the bias supply characteristics.

If that was not enough to contend with, it can also be seen that the bias supply negative DC voltage is permanently in the signal circuit.

Furthermore, the bias voltage is offset negatively to the centre-line axis of the signal voltage, which is nominally at ground potential. The greater the applied bias voltage the greater the offset.

Since the AC signal alternations in both halves of the push-pull circuit have both positive and negative polarity, it follows that the bias supply voltage will enhance the signal in the negative alternation and offset the signal in the positive alternation.

This DC voltage and current source will try to support the flow of AC signal sourced energy to the grids when grid current flows in Classes AB1 and AB2 or B - and also Class A if gas is present in the power tubes . However to get to the grids it must first pass through the driving tube circuit. Since the interstage coupling capacitor will not pass DC it follows that the DC component of grid drive power will be dissipated into the plate resistor of the driving tube.

But along the way it can increase the cathode bias voltage of the driving tube (which will attenuate the signal) whilst simultaneously reducing the plate voltage (which will attenuate the signal and reduce AC output voltage)  - thereby affecting the capability of the driving stage to respond to the signal input.

Since the plate load resistor of the driving tube provides an alternative return circuit (to ground) for the signal voltage it follows that the conventional fixed bias configuration which adds DC currents into the signal circuit includes elements guaranteeing circuit instability. When negative loop feedback is added to such a circuit - eg from the speaker secondary windings of the output transformer back to the driving stage - then the result must be difficult to predict or control.

Finally, the bias supply final shunt load resistor is in series with the output tube grid resistor and forms an integral part of the grid to cathode circuit of the output tubes. It is common to see relatively high values of bias supply final shunt load resistor compared with the output tube grid resistor value - eg 47k to 100k. Sometimes complex bias adjusting networks are also in there. It follows that the closer the (total) bias supply resistor is to the grid resistor value then the more effect the bias supply characteristics will have upon the AC signal behaviour.


Fig 5 shows the symmetrical balanced AC signal drive system whereby the AC signal OUTPUT from the Driving stage circuit and the AC signal INPUT of the Driven stage circuit have a common central axis at reference voltage potential.

In this example, the bias is set at -40 VDC.

Notice how using this system as shown in Fig 4, the signal path circuits on both sides of the push-pull circuit are exactly equal and balanced - in a similar manner to that of Fig 1: Cathode Bias Output Stage.

Moreover, the signal path is direct between both driving and driven stages and is completely independent of and therefore unaffected by the bias supply.

Consequently neither actual bias voltage nor bias supply characteristics will affect the signal circuit.

Note that the grid to cathode circuits of the power tubes is the same for both Fig 5 and Fig 6 designs.

This system is very stable and provides enhanced frequency response and reduced distortion.

It is entirely suitable for a cathode-follower driver stage - ensure the base of the cathode follower load resistor is connected to the bias supply as shown in Fig 5.

An added benefit for a cathode-follower driver system is that the cathode load resistor now spans the bias voltage as well as the B+ voltage. Consequently the cathode follower load resistor may be made substantially larger whilst still maintaining the same plate to cathode voltage in the tube. This can offer increased output voltage to the driven stage.

Load Resistor

Generally speaking, the greater the load resistance/impedance to the driving stage the higher the output capability.

However there are practical limits to this objective since output tube grid circuit resistance must be held within manufacturers' published specifications.

To maximise the driver stage load whilst minimising driven stage grid circuit resistance the RATIO of driver stage load plate resistor to driven stage grid circuit load resistor should be held within the range of about 1:1 to 1:2.


Most of us have a junk box stock of perfectly good tetrodes, pentodes or beam power tubes just waiting to be used - so why not experiment and prove the benefits of OPTIMISED ELECTRON STREAM TECHNOLOGY to yourself.


There is no restriction or cost imposition upon the home hobbyist constructor to using these concepts - the only restriction is on commercial exploitation - so if you do not like it do not do it.

If you want your hi-fi to improve its performance at minimal cost to you then experiment. The concepts presented here do work and cost very little to implement.

However to those who say that a product is only as good as what you pay for it, then these concepts are of no value to you because they are free. You would be wiser to spend a hundred grand on a commercial system and feel better. While you are so doing, ask the manufacturer to justify the circuit design and component choices to you - ie why the design is what it is and not some other alternative approach. That is "why is it better?"

To those who consider relating these concepts to RF applications - this is an audio focused site - experiment at your own risk. It may be safer to stay with the tried and true - waste a little power and live with it.

However RCA, in their Transmitting Tube Manual TT4, say at Page 44 -" The only restrictions on tube operating values are those imposed by the published maximum ratings." The OPTIMISED ELECTRON STREAM TECHNOLOGY design concepts presented above should enable designers to withdraw operating conditions back into specification whilst improving performance.

Finally, an OES tube amplifier can only and will always sound like a tube amplifier. The OES concept is limited to and by the vacuum tubes it relates to.












Intellectual property in the applied engineering concepts expressed in this paper remains exclusively with the author.







For suggestions, critique or discussion re this page contact:


Email:   contact

This page is located at http://www.oestex.com/tubes/oes.html

OES Home Page

This page last amended 29 December 2017